Lascar Electronics

<p>A major challenge in any circuit emulation service (CES) applications is clock recovery at the far end of the link. An interworking function (IWF) supports differential and adaptive clock recovery. But for more demanding applications, such as synchronization of wireless base stations, the hybrid timing generator can be applied.</p>

Making an Input/Output Phase measurement The other common phase measurement compares the phase of the signal at the input of the DUT to the phase of the same signal at its output. A simple way to make this measurement for the Left channel in our DUT is to select GenMon as the Channel A analyzer input and then connect the left channel DUT output to the Channel B analyzer input.

We have so far considered only unimpeded straight-line propagation, and specular reflection (where the angle of incidence and the angle of reflection are the same). In UHF RFID, the typical wavelengths of around 32 cm are comparable to the size of many obstacles present in the environment, so to fully treat the propagation environment, we must account not only for propagation and reflection, but diffraction: the ability of obstacles of finite size to scatter the incident radiation in directions other than specular. The full treatment of diffraction is rather complex, and not as important for passive RFID as for other communications fields since the forward link budget is so small.

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The importance of diffraction may be roughly estimated by calculating the effective size of an obstacle in terms of the phase difference between the shortest and longest paths through the obstacle (Figure 3.39). In the figure, the shortest distance is the direct path (which passes right through the obstacle) and the longest distance goes around the edge of the obstacle. The phase difference is the difference in these path lengths multiplied by the wavenumber k = ω / c:

(A typical value of k for UHF RFID is about 19-20 radians/m.) This difference, measured in half-wavelengths (i.e. δ φ/ π, since there are 2π radians of phase in one wavelength), is the number of Fresnel zones subtended by the object. When this number is small (on the order of 1-2 or less), diffraction is important, and the received intensity is a complex function of position, with no well-defined shadow region. When the obstacle subtends many Fresnel zones (>3-5), it is able to form a fairly well-defined shadow, and tags in that region are unlikely to be visible to the reader.

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Consider, for example, a disk of diameter 1 m, illuminated by a reader antenna 2 m from the disk. If a tag is placed 5 cm behind the disk, the difference between the direct path and the path that goes around the edge of the disk is about (2.56″2.05)=0.51 m, which is about 3 Fresnel zones at 915 MHz (0.51/0.16). The tag is likely to find itself in a deep shadow and not be read (though it is worth noting that if the tag is carefully positioned exactly along the axis of the disk, it will find itself in a relatively high-intensity region in the middle of the shadow, known as Poisson's bright spot, which may allow it to power up). On the other hand, if the tag is placed 2 m from the disk, the path length difference becomes 0.12 m, rather less than one Fresnel zone, corresponding to weak and complex shadowing.

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In this position, the tag will move in and out of faded regions as its position relative to the disk and reader changes. In practice, such weak shadowing often simply adds to the complex fading behavior resulting from walls, floors, and other obstacles that can be treated as specular reflectors. Thus, obstacles that are small relative to a wavelength, or distant from both the transmitter and the receiver (reader antenna and tag) have modest though nonnegligible effects, and tags may be read even though the straight-line path from reader to tag is obstructed.

Obstacles that are large compared to a wavelength, and close to either the reader antenna or the tag, are likely to prevent passive tags from being read. It is in this sense that RFID is a non-line-of-sight technology even for metallic obstacles.

Figure 7 has the same load and ripple current as before. Here, Vout is set to 2/3 Vin . Without any slope compensation in this example, we see that perturbed inductor current is unable to converge to the desired steady state pattern and its variation in PWM is even more erratic than in the 50 percent example. The math says that this waveform will eventually settle to a sub-harmonic oscillation as well.

Adding slope compensation The next series of plots (Figs. 8-10) show the same power converter applications (fixed Vin , Vout, L, and load) with the addition of an artificial slope compensation signal, I-slope , to the summing node of the CMC circuit. How does this additional time dependent bias signal help avoid the runaway condition seen in Figs. 6 and 7?

Notice that without slope compensation for duty cycles greater than 50 percent, Sd has a greater magnitude than Su . This means that for perturbed currents, and for higher duty cycles, the downward current moves away from the control signal at a greater rate. The rate of divergence of the downward inductor current from the control signal becomes greater than the rate of convergence of the upward inductor current to the control current in this region of high duty cycle. Intuitively, the higher rate of divergence during the 'off' time leads to overall divergence of the inductor current and therefore instability.

Backward compatibility, for instance, is a requirement.IEEE802.3at-compliant PDs will recognize both IEE802.3af and IEEE802.3at-compliant PSEs. At the same time 802.3at-compliant PSEs will recognize PDs compliant with either standard. Also, IEEE802.3af-compliant PSEs will recognize new Class 4 IEEE802.3at-compliant PDs (Class 4 was included in the IEEE802.3af standard but reserved for future use).

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